1. Field of the Invention
The present invention relates to a pilot interpolation synchronous detection method for a transmission circuit in a radio communication system and, more particularly, to an interpolation synchronous detection method and radio communication system which can be used for, for example, a pilot interpolation synchronous detection spread spectrum scheme.
2. Description of the Prior Art
Recently, a pilot interpolation synchronous detection spread spectrum scheme has been proposed as one of the mobile communication schemes in RCS 94-98 “Characteristics of Interpolation Synchronous Detection RAKE in DS-CDMA” by ARIB (Association of Radio Industries and Businesses).
In pilot interpolation synchronous detection, first and second pilot signals whose phase points are known are cyclically or periodically inserted in an information signal to form a frame, and a transmission path that varies due to multipath Rayleigh fading is estimated in the interval between the first and second known pilot signals. Letting Z1 and Z2 be the coefficients (transfer functions) estimated from the first and second known pilot signals, a coefficient (transfer function) Z(k) which is obtained by estimating a transmission path at the kth symbol of N symbols of the information signal can be obtained by primary interpolation of coefficients Z1 and Z2 of the first and second known signals as per:Z(k)=[(N−k)/N]Z1;[k/N]Z2  (1)
Furthermore, kth demodulated data Sk obtained upon pilot interpolation synchronous detection is the product sum from i=1 to i=p expressed as:Sk=[αi×Z*i,k*×ri,k]  (2)where p is the number of delayed waves to be subjected to RAKE reception, αi is the weighting coefficient for the ith delayed wave, Z*i,k* is the complex conjugate of the coefficient phase estimated and primarily interpolated by interpolating the ith delayed wave estimated on the basis of the coefficients Z1 and Z2 estimated with respect to the ith delayed wave, and ri,k is the signal obtained by despreading each reception signal of the ith delayed wave.
When the multipath influences can be isolated from each other by despreading in this manner, interpolation synchronous detection using known symbols can be performed for each despread signal before RAKE synthesis, as indicated by equation (2).
If the multipath delay difference is larger than ±1 chip which is a delay difference allowing isolation of multipath influences by despreading, synchronous detection can be performed for each transmission path by interpolation synchronous detection. If, however, the multipath delay difference is smaller than ±1 chip which is the minimum difference allowing isolation of multipath influences by despreading, it is difficult to perform interpolation synchronous detection using known symbols for each despread symbol for the following reason. Even if the delay difference is small, the influences of different transmission paths are independent. Basically, therefore, transmission path estimation must be performed independently.
In practice, multipath signals having a delay difference within ±1 chip are received on the receiving side with intersymbol interference, and it is generally difficult to remove the influences of the interference as in general radio communication schemes, other than the spread spectrum communication scheme, in which the influences of transmission path distortion due to multipath transmission cannot be removed.
For this reason, when despreading and interpolation synchronous detection are to be performed by selecting one optimal reception sampling point (e.g., a sampling point at which the eye pattern of a reception signal opens most) in predetermined cycles from the reception signals oversampled at n points (four points a to d in the following case), if there are transmission paths exhibiting a delay difference within ±1 chip, the reception sampling point to be selected changes. In addition, since the influences of transmission paths at the respective reception sampling points can be regarded as independent, interpolation synchronous detection may not be properly performed.
Furthermore, in a radio communication system, the reception signal power dynamic range is generally very large. A reception section in a terminal radio communication unit, in particular, uses a method of realizing a large dynamic range by combining a gain control section whose gain changes stepwise and a gain control section whose gain continuously changes. In this case, the gain of the overall reception section can be continuously changed in a wide range by a kind of amplitude range switching operation.
When the gain is continuously changed, the phase rotation amount of a reception signal in a receiver may undergo a discontinuous change at a point at which range switching is performed by switching the gain control section whose gain changes stepwise. This may make it impossible to perform normal interpolation synchronous detection as in the case described above in which despreading and interpolation synchronous detection are performed by selecting one optimal reception sampling point (e.g., a sampling point at which the eye pattern of a reception signal opens most) in predetermined cycles from the reception signals oversampled at n points. The above description has been made by taking gain changes in the reception section as an example. Obviously, however, this applies to the transmission section on the other party.
The above problems will be described in detail below with reference to FIGS. 1A to 1C. FIG. 1A shows a frame configuration of reception signal frames each containing a pilot symbol for interpolation synchronous detection and the timing of oversampling. In this case, quadrature oversampling is performed at points a, b, c, and d. FIG. 1B shows the timing at which an optimal sampling point for demodulation (e.g., a sampling point at which the reception eye pattern opens most) is selected from the points at which quadrature oversampling is performed. FIG. 1C shows the transition of a reference phase point with respect to each sampling point. Referring to FIG. 1C, a straight line passing through points q and s represents a reference phase transition at the sampling point c, and a straight light passing through points r and t represents a reference phase transition at the sampling point b. A difference Φ between these straight lines with respect to the ordinate (phase) represents the relative phase difference between a path that reaches the reception section at the timing corresponding to the sampling point b and a path that reaches the reception section at the timing corresponding to the sampling point c.
Assume that the optimal sampling timing for demodulation changes from b to c to b. In this case, in the prior art, as shown in FIG. 1B, since the sampling timing is updated immediately before (or after) a pilot symbol, the reference phases measured at the respective update timings are represented by p, q, and t.
In addition, since interpolation synchronous detection is performed, the transition of an estimated reference phase between pilot symbols is expressed by line segments p-q and q-t. In this case, the phase transition at the actual sampling point is represented by line segments p-r and q-s. Consequently, the integral value of estimated reference phase errors can be calculated from the areas of triangles prq and qst, each of which is given by(1/2)·Φ·LTherefore, this area matches with an error component, and an error in a linearly interpolated estimated transfer function increases, resulting in a deterioration in the accuracy of demodulated data.
The above description is about the spread spectrum communication scheme. However, this applies to an interpolation synchronous communication system other than the spread spectrum communication scheme except for despreading processing.